This invention relates to DC-to-DC voltage converters and, more particularly, to DC-to-DC boost converters which minimize switching losses in converter semiconductor devices by using zero voltage switching techniques.
A DC-to-DC boost converter is usually chosen as the front end power stage in an AC/DC converter. DC-to-DC converters can be of two types: pulse width modulated (PWM) converters and resonant converters. PWM converters interrupt power flow and control the duty cycle to process power. Resonant converters process power in sinusoidal form. PWM converters operate at a constant frequency and variable pulse width, while resonant converters operate with variable frequency at a constant pulse width. PWM converters are used predominantly today because of circuit simplicity and ease of control.
In a PWM boost converter circuit a switch is rapidly switched on to create a high voltage across an inductor. When the switch is turned off, the inductor current charges an output capacitor through a diode and creates a voltage at the output higher than the original supply voltage. FIG. 1 illustrates a basic (PWM) boost converter circuit 100 comprised of a MOSFET power transistor (MOSFET) 102, inductor 104, diode 106 and a capacitor 108. The gate terminal "g" of the MOSFET 102 is connected to an external pulsed switching voltage source (Vswitch) 116. The drain terminal "d" of the MOSFET 102 is connected to the inductor 104 and the diode 106. The source terminal "s" of the MOSFET 102 is connected to ground. FIG. 1 shows a voltage source (Vin) 112 to be connected at the input to the circuit 100, and a load 114 connected in parallel to the capacitor 108 at the output of the circuit 100. FIG. 1 also shows reference currents ILl04 through the inductor 104, ID106 through the diode 106, and IDS102 through the MOSFET 102; and reference voltages VL104 across inductor 104, VDS102 across MOSFET 102, VC108 across capacitor 108, and VL across the load 114.
MOSFET 102 functions as an electronic switch to control the current IL104 through the inductor 104. During the converter switching cycle a pulsed voltage is applied to the gate of the MOSFET 102 by Vswitch 116. This pulsed voltage cycles the MOSFET 102 from the "on" (conducting) to the "off" (nonconducting) state. When MOSFET 102 is on and conducting, the drain-source voltage VDS102 across MOSFET 102 is zero and the current ILl04 is driven from Vin through inductor 104 and MOSFET 102 to ground. The currents IL104 in inductor 104 and IDS102 in MOSFET (102) are equal at this time. During this stage of the switching cycle VL is supplied to the load by voltage VC108 on capacitor 108 which was charged in the previous cycle. Diode 106 blocks reverse current flow from the capacitor 108 into MOSFET 102 and ground.
When MOSFET 102 is switched off, the interruption in the flow of current IDS102 through MOSFET 102 creates a higher voltage across the inductor 104. At switch off, the voltage VL104 across inductor 104 instantaneously changes polarity and rises to the difference between Vin and VL. The Diode 106 is now forward biased and the energy stored in inductor 104 is discharged into capacitor 108 and the load 114 by the current ID106 through diode 106. The current through inductor 104 decreases and the voltage VC108 on capacitor 108 increases.
The previous cycle of switching MOSFET 102 on and off is repeated. After a set time MOSFET 102 is again turned on. The converter is automatically controlled so that the average current in inductor 104 equals the load current. Current is again driven from Vin to ground through inductor 104 and MOSFET 102, while capacitor 108 supplies the load 114 with charged energy stored during the previous cycle. The average voltage on capacitor 108 depends on the pulse-width of the output of Vswitch 116. The cycle of switching MOSFET 102 between on and off is repeated at a very high rate. The pulsed voltage applied by Vswitch 116 can be typically at a frequency of 30-50 kHz. A high converter switching frequency is desirable because higher frequencies allow the use of inductors smaller in both value and size. The converter can then be packaged in a smaller volume and will be lighter in weight.
One drawback of operating a converter at high switching frequencies, however, is the switching power losses which increase as the switching frequency increases. In practice, these switching losses are the limiting factor for the choice of switching frequency. One goal of converter design has been to operate at high frequencies while minimizing switching losses in the switching elements of the converter.
In the boost circuit 100 of FIG. 1 losses occur both during turn-on and turn-off of MOSFET 102. A MOSFET, such as MOSFET 102, has an internal parasitic capacitance which is effectively a capacitance across the drain-source terminals. This drain-source capacitance causes MOSFET 102 to inductively turn off and capacitively turn on. During turn-off, voltage spikes induced by the leakage inductances cause noise and voltage stress. During turn on, the energy stored in MOSFET 102's drain-source capacitance is dissipated internally. The turn-on loss depends on the switching frequency and the energy stored in the drain-source capacitance.
Another cause of switching losses in the boost circuit 100 are turn-on losses in the switching transistor MOSFET 102 due to reverse recovery current in the diode 106 before diode 106 turns off. When MOSFET 102 is turned on there is a finite time required for recombination of charges in diode 106. Until these charges in diode 106 recombine, a negative spike of reverse recovery current is generated in diode 106. The energy from this current spike is dissipated in MOSFET 102.
The turn-on switching losses from the MOSFET 102 drain-source capacitance and the diode 106 reverse recovery current are shown in FIGS. 2A-D which illustrate current and voltage waveforms for the MOSFET 102 turn-on portion of the switching cycle. FIG. 2A shows VDS102, the drain-source voltage on MOSFET 102. FIG. 2B shows ID106, the current through diode 106. FIG. 2C shows IDS102, the drain-source current through MOSFET 102. FIG. 2D shows Vswitch, the applied gate-source voltage on MOSFET 102. As may be seen from FIGS. 2A-D, the MOSFET 102 turn on portion of the switching cycle may be divided into five ti me interval s I-V.
During interval I, Vswitch is zero and MOSFET 102 is off. VDS102 is at the output voltage level plus the voltage drop across diode 106. At the beginning of interval II, Vswitch begins to rise as it is pulsed high to turn MOSFET 102 on. During Interval II Vswitch is below the threshold voltage needed to turn MOSFET 102 on. During interval III MOSFET 102 turns on as Vswitch rises above the turn-on threshold. VDS102 decrease by as the drain-source capacitance of MOSFET 102 discharges, and diode 106 becomes reverse biased and begins to turn off. The current ID106 goes negative because of the large pulse of reverse recovery current in diode 106. Since there is no limiting resistor in series with MOSFET 102 and diode 106, the current ID106 is relatively large. Because the voltage VDS102 is still high, large power losses occur in MOSFET 102. In interval IV diode 106 has turned off. More losses occur in MOSFET 102 as VDS102 collapses to zero. In interval V, Vswitch rises and MOSFET 102 becomes saturated and fully turned on.
The losses in MOSFET 102 during intervals III and IV of the turn on cycle can be limited by minimizing the voltage VDS102 across MOSFET 102 and the reverse recovery current ID106 in diode 106, which flows into MOSFET 102 during the turn-on portion of the switching cycle. The ideal turn on condition would be to set VDS102 across MOSFET 102 at zero volts. With VDS102 at zero volts during turn on, the product of voltage times the current in MOSFET 102 and, therefore the power dissipation, would be zero. This goal can be realized by applying known zero voltage switching techniques to the basic boost converter circuit of FIG. 1.
Zero voltage switching is a technique whereby the drain-source capacitance of MOSFET 102 is caused to discharge in a quasi-sine wave fashion so that the device can be switched at the instant the voltage across it is at zero. Traditionally, zero voltage switching techniques have been used to transform in PWM converters into hybrids of PWM and resonant converters. These hybrids are known as quasi-resonant converters. While these quasi-resonant converters minimize switching power losses, they do not operate as true PWM converters. In a quasi-resonant converter the voltage across the switching device can be up to twice the output voltage. Thus, Quasi-resonant converters require a switching device which can withstand a voltage over twice the output voltage in contrast to PWM converters which require a switching device which can withstand just the output voltage. Quasi-resonant converts also operate at a variable frequency. It would be desirable, however, to have a PWM boost converter which operates at a constant frequency, in continuous mode (at a constant current), and which has minimal switching losses.
The present invention provides a PWM converter design allowing for zero voltage switching at turn-on and minimal switching losses. This PWM boost converter design operates as a true PWM converter. Furthermore, it can be operated at higher frequencies and is smaller in size and weight. Also, because it operates at a constant frequency, simpler input and output filters may be used with the converter circuit of the present invention.